1. Field of the Invention
This invention relates to switching converters particularly adapted to switch at relatively high frequencies and, in particular, to such converters that achieve switching on and off at zero current level, whereby high efficiency at such high frequencies is achieved.
2. Description of the Prior Art
In conventional pulse width modulation (PWM) switching DC-to-DC converters, a switching device typically in the form of a semiconductor switch turns on and off repetitively at high current levels to achieve output voltage conversion and regulation. Such converters employ magnetic components and capacitors for energy storage/transfer and ripple/filtering. Operating such magnetic components and capacitors at high frequencies reduces their size and cost. In typical PWM converters, the inductive impedance of such magnetic components is coupled in circuit with the semiconductor switches. High frequency switching of such inductive impedances, adversely affects these switches. As the switch is turned on and off rapidly, switching transients involving high levels of current and voltage occur, whereby high switching stresses and losses are imposed upon the semiconductor switch. When such a switch is switched or "forced off", the energy still present in the coupled inductive element imposes high current and high voltage and thus high switching stress and loss on the switch. Furthermore, the pulsating current waveforms resulting from rapid switching, cause severe electromagnetic interaction (EMI) problems as the switching frequency is increased. It is desired to switch such semiconductor switches at relatively high switching frequencies to increase the effectiveness of the voltage control and regulation and, at the same time, minimize the size and cost of the inductive and capacitive elements employed in such converters. However, as the switching frequency increases, the above-noted switching stresses and losses increase and the converter's overall efficiency and reliability decrease.
Snubber circuits are commonly used to alleviate the switching stresses mentioned above. Simple RC or RDC snubber circuits suffer from high power loss at high frequencies. Lossless snubber circuits, on the other hand, increase circuit complexity considerably.
To overcome these problems of switching stress and loss, the technique of "zero current switching" has been described in "Resonant Switching Power Conversion Technique," by E. E. Buchanan and E. J. Miller, IEEE Power Electronics Specialists Conference, 1975 Record, pp. 188-193 and in "Resonant Switching Power Conversions," by E. J. Miller, IEEE Power Electronics Specialists Conferences, 1976 Record, pp. 206-211. Such "zero current switching" technique utilizes an LC resonant tank circuit to force the current through the semiconductor switch to oscillate, whereby the semiconductor switch turns off at zero current level, thereby drastically reducing switching stresses and losses.
To generalize the zero-current switching technique, the concept of resonant switch was described in "Resonant Switches--A Unified Approach to Improve Performance of Switching Converters," by the inventors of this invention, IEEE International Telecommunications Energy Conference, 1984 Proceedings, pp. 344-351. This paper shows the use of "resonant switches" in various conventional pulse-width modulated switching converters to achieve "zero-current-switching". Generally, such resonant switches are a subcircuit consisting of a semiconductor switch S.sub.1, a resonance inductor L.sub.r, and a resonance capacitor C.sub.r. There are two types of resonant switch configurations as shown respectively in FIGS. 1A and B, an L-type and an M-type resonant switch. In both cases, the inductor L.sub.r is connected in series with the switch S.sub.1 to slow down the current change rate, and the capacitor C.sub.r is added as an auxiliary energy storage/transfer element. If switch S.sub.1 is a device without reverse voltage blocking capability or contains an internal anti-parallel diode, an additional diode D.sub.1 is needed and should be connected in series with the switch S.sub.1 and the inductor L.sub.r, as shown in FIGS. 1C and 1D. The inductor L.sub.r and the capacitor C.sub.r together constitute a series resonant circuit with respect to the switch S.sub.1. When the switch S.sub.1 conducts, current flows through switch S.sub.1 and inductor L.sub.r into the capacitor C.sub.r with a quasi-sinusoidal waveform. As the inductor current drops to zero, the capacitor voltage is charged up with a polarity that reverse biases the switch S.sub.1, thus commutating off the switch S.sub.1. The resonant switch therefore, provides zero-current-switching properties during both turn on and turn off.
A conventional buck converter is illustrated in FIG. 2A, as comprising a switch S.sub.1 for applying upon being rendered conductive a voltage source V.sub.s across a commutation diode D. The commutation diode D is coupled to an output circuit comprised of an output inductor L.sub.o disposed in circuit with an output capacitor C.sub.o connected in parallel with an output resistor R.sub.o. This conventional buck converter is modified as shown in FIG. 2B by the addition of the L-type resonant switch, as first shown in FIG. 1A, between voltage source V.sub.s and the commutation diode D. The output inductance L.sub.o is selected to be much larger than inductance L.sub.r, thus making the resonant frequency of the resonant circuit comprised of capacitor C.sub.o and the inductor L.sub.o much smaller than that of the resonant circuit comprised of the capacitor of C.sub.r and the resonant inductor L.sub.r. It is also assumed that inductor L.sub.o is sufficiently large so that the current I.sub.2 through the inductor L.sub.o, remains relatively constant throughout a switching cycle.
The operation of the buck resonant converter employing the L-type resonance switch as shown in FIG. 2B, will now be explained with reference to the waveforms as shown in FIGS. 3A to 3D. Before time T.sub.0, the semiconductor switch S.sub.1 is turned off, whereby the commutation diode D carries the output current I.sub.o with the capacitor voltage V.sub.c clamped at zero. In the first of four distinct switching stages, the semiconductor switch S.sub.1 is turned on at time T.sub.0, whereby current I.sub.1 flowing through the semiconductor switch S.sub.1 and the resonant inductor L.sub.r rises linearly as shown in the waveform of FIG. 3B. Between times T.sub.0 and T.sub.1, the output current I.sub.2 shifts gradually from the path through the commutation diode D to the path through the semiconductor switch S.sub.1 and the resonant inductor L.sub.r l.
At time T.sub.1, the current I.sub.1 becomes equal to current I.sub.2, whereby the commutation diode D is turned off and, as seen in FIG. 3B, the current I.sub.1 l begins to charge capacitor C.sub.r. As seen in FIG. 3B, the flow of the current of I.sub.1 through the resonant inductance L.sub.r and the voltage V.sub.c appearing on resonant capacitor C.sub.r is substantially sinusoidal rising to a peak and falling back to zero at time T.sub.2. As shown in FIG. 3D, the voltage V.sub.C across the resonant capacitor rises to a peak of approximately 2V.sub.s shortly before time T.sub.2, whereby a reverse voltage of V.sub.c -V.sub.s is applied to the semiconductor switch S.sub.1 commutating it off naturally at time T.sub.2. As shown in FIG. 3B, zero current is flowing in the semiconductor switch S.sub.1 at time T.sub.2, when it is commutated off. As shown in FIG. 3D, the capacitor C.sub.r discharges in the time interval from time T.sub.2 to time T.sub.3. The capacitor voltage V.sub.c drops linearly to zero at time T.sub.3. In the fourth stage from time T.sub.3 to time T.sub.4, the output current I.sub.2 flows through the commutation diode D and, with the switch S.sub.1 open, the resonant capacitor C.sub.r is clamped to zero voltage. At time T.sub.4, the switch S.sub.1 turns on again, starting the next switching cycle.
FIG. 2C shows a buck resonant converter circuit in which the resonant capacitor C.sub.r is coupled in parallel between the voltage source V.sub.s and the resonant inductor L.sub.r instead of in parallel with the commutation diode, whereby an M-type resonant switch, as shown first in FIG. 1B, is formed. The modified buck resonant converter of FIG. 2C operates in four stages in a manner similar to the operation of the buck resonant converter as described above with respect to FIG. 2B.
The operation of the converter circuits with the L-type and M-type resonant switches as shown in FIGS. 2B and 2C, is in the half-wave mode as shown in FIG. 3B. In other words, the current I.sub.1 is permitted to flow through the switch S.sub.1 in but a single direction. As will be explained below, these resonant converters as operated in the half-wave mode suffer from a draw back, namely, the DC voltage conversion ratio is sensitive to load variations.